Digital timing extraction and recovery in a digital video decoder

ABSTRACT

In one example, a digital video resampler and decoder is disclosed. An input buffer is coupled to a source of a video signal oversampled by at least two times. A horizontal synchronization detector is coupled to the input buffer, and detects horizontal synchronization boundaries. A counter is coupled to the horizontal synchronization detector to count a number of samples between horizontal synchronization boundaries. A comparator is coupled to the counter and compares the counted number of samples to a reference count. A sample corrector is coupled to the input buffer and modifies a block of samples based on a result from the comparator. An output buffer is coupled to the sample corrector to hold the modified block of samples. A comb filter is coupled to the output buffer, and generates first and second three-dimensional color values based on the modified block of samples.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims the benefit under 35 U.S.C. 119(e) of U.S.Provisional Application Ser. No. 61/020,718, filed Jan. 12, 2008, whichis incorporated herein by reference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The invention relates to digital video decoders and in particular todigital timing extraction and recovery in a digital video decoder.

2. Background of the Invention

Due to advancing semiconductor processing technology, integratedcircuits (ICs) have greatly increased in functionality and complexity.With increasing processing and memory capabilities, many formerly analogtasks are being performed digitally. For example, images, audio and fullmotion video can now be produced, distributed, and used in digitalformats.

Although digital images generally provide higher noise immunity, mostdigital images in digital video streams are converted from analog videostreams. The original analog video stream may contain noise from varioussources. For example, modulation, wireless transmission and demodulationof TV signals may introduce Gaussian-like noise. Furthermore, evenanalog video transferred over transmission lines may have Gaussian-likenoise due to magnetic fields around the transmission lines. In addition,the digitalizing process may inadvertently amplify minor noise problemsin the analog video stream. For more information on methods of noisereduction for interlaced digital video stream, see United States PatentPublication 20070103594 (application Ser. No. 11/644,855 by Zhu,published on May 10, 2007, filed on Dec. 22, 2006, titled “Recursivenoise reduction with still pixel detection” and assigned to HuayaMicroelectronics, Ltd.), the contents of which are incorporated hereinby reference.

FIG. 1A is an illustrative diagram of a portion of interlaced digitalvideo signal 100′ most often used in television systems. Interlaceddigital video signal 100′ comprises a series of individual fields F(0)to F(N). Even fields contain even numbered rows while odd fields containodd numbered rows. For example if a frame has 400 rows of 640 pixels,the even field would contains rows 2, 4, . . . 400 and the odd fieldwould contains rows 1, 3, 5, . . . 399 of the frame. In general, for aninterlaced video signal each field is formed at a different time. Forexample, an interlaced video capture device (e.g. a video camera)captures and stores the odd scan lines of a scene at time T as fieldF(5), then the video capture device stores the even scan lines of ascene at time T+1 as field F(6). The process continues for each field.Two main interlaced video standards are used. The PAL (Phase AlternatingLine) standard, which is used in Europe, displays 50 fields per seconds(fps) and the NTSC (National Television System Committee) standard,which is used in the United States, displays 60 fps. Interlaced videosystems were designed when bandwidth limitations precluded progressive(i.e., non-interlaced) video systems with adequate frame rates.Specifically, interlacing two 25 fps fields achieved an effective 50frame per second frame rate because the phosphors used in televisionsets would remain “lit” while the second field is drawn.

To ease transmission of video signals, chrominance information andluminance information are combined via modulation into a singlecomposite video signal. Imperfect decoding of composite video signals ineither PAL or NTSC format may lead to color-crossing. Specifically,color-crossing error often appears in a video image where the localluminance spatial frequency is near the sub-carrier frequency of thechrominance information. Color-crossing errors occur in both PAL andNTSC video signals.

For example, NTSC video signals typically have a chrominance sub-carrierfrequency of 3.58 MHz, i.e., chrominance information is modulated by asinusoid signal with a frequency equal to 3.58 MHz before transmission.Luminance information may also have components that overlap with thechrominance information near the chrominance sub-carrier frequency.Thus, the luminance components near the chrominance sub-carrierfrequency cause color-crossing errors, which cannot be cleanly removed.Generally, during video decoding a band pass filter at the chrominancesub-carrier frequency is used to obtain the chrominance information.However, the luminance components, which are near the chrominancesub-carrier frequency, are not blocked by the band pass filter.Therefore, the decoded chrominance signal would include “unclean”chrominance information. The color-crossing errors produce rainbow likecolor blinking in the decoded video image. In PAL video signals, thesame color-crossing errors also occur at the PAL chrominance sub-carrierfrequency of 4.43 MHz. Color-crossing error can also occur in otherencoded video signals.

Conventionally, 3D comb filters have been used to reduce color-crossingerrors. Specifically, in NTSC composite video signals the chrominance ofcorresponding pixels in two consecutive fields of the same type (odd oreven) have a phase difference equal to 180 degrees. A 3D comb filter cancancel the miss-included luminance components by a simple subtraction ofthe video signal values of the two corresponding pixels, when the videoimage is not changing. However, for PAL composite video, the chrominanceof corresponding pixels in two consecutive fields of the same type haveonly a 90-degree phase difference. Thus, to use 3D comb filters tocorrect color-crossing errors in decoded PAL composite video signals,four fields must be used.

While 3D comb filters can reduce color-crossing errors, 3D comb filtersmay also degrade other aspects of video quality. For example, 3D combfilters are very sensitive to noise in composite video signals;therefore, a digital video decoder with a 3D comb filter would havedifficulties with weak video signals, which are common in many areas.Furthermore, high quality 3D comb filters are very expensive relative toother components of a video system. For more information on efficientreduction of color-crossing errors from decoded composite video signals,see United States Patent Publication 20060092332 (application Ser. No.11/046,591 by Zhu, published on May 4, 2006, filed on Jan. 28, 2005,titled “Color-crossing error suppression system and method for decodedcomposite video signals” and assigned to Huaya Microelectronics, Ltd.),the contents of which are incorporated herein by reference.

Modern video signals typically consist of a sequence of still images, orframes or fields as described above. By displaying the sequence ofimages in rapid succession on a display unit such as a computer monitoror television, an illusion of full motion video can be produced. Astandard NTSC television display has a frame rate of 29.970 fps (framesper second). For historical reasons, the frames in video displays formost consumer applications (and many professional applications) areformed from “interlaced” video signals in which the video signals aremade up of “fields” that include half the data required for a fullframe. As described above, each field includes every other row of pixelsthat would be included in a complete frame, with one field (the “oddfield”) including all the odd rows of the frame, and the other field(the “even field”) including all of the even rows.

FIG. 1B depicts this interlacing concept, as a view 110′ is interlacedinto an odd field 120′ and an even field 130′. Odd field 120′ includesodd rows SO(1), SO(2), SO(3), SO(4), SO(5), SO(6), SO(7), and SO(8),which represent rows 1, 3, 5, 7, 9, 11, 13, and 15, respectively, ofview 110′. Even field 130′ includes even rows SE(1), SE(2), SE(3),SE(4), SE(5), SE(6), SE(7), SE(8), which represent rows 2, 4, 6, 8, 10,12, 14, and 16, respectively, of view 110′. Note that each of odd rowsSO(1) SO(8) in field 120′ corresponds to a blank row (i.e., a row withno pixel values) in field 130′, while each of even rows SE(1) SE(8) infield 130′ corresponds to a blank row in field 120′.

View 110′ depicts a white square 111′ formed in a shaded background112′. Therefore, odd rows SO(1) SO(8) are all shaded, except for a whiteportion 121′ in each of odd rows SO(4), SO(5), and SO(6) correspondingto the portion of those rows corresponding to white square 111′.Similarly, even rows SE(1) SE(8) are all shaded, except for a whiteportion 131′ in each of even rows SE(3), SE(4), and SE(5), correspondingto the portion of those rows corresponding to white square 111′.

Note that color video signals contain chrominance and luminanceinformation. Chrominance is that portion of video that corresponds tocolor values and includes information about hue and saturation. Colorvideo signals may be expressed in terms of RGB components: a redcomponent R, a green component G, and a blue component B. Luminance isthat portion of video corresponding to brightness value. In a black andwhite video signal, luminance is the grayscale brightness value of theblack and white signal. In a color video signal, luminance can beconverted into red, green and blue components, or can be approximated bya weighted average of the red, green and blue components. For example,in one well-known scheme, luminance is approximated by the equation:Y=0.30*R+0.59*G+0.11*B. For explanatory purposes, shaded regions of thefigures represent lower luminance values than blank (white) regions. Forexample, the white portion 121′ in odd row SO(4) has a higher luminancevalue than the shaded portion of the same row.

To generate a progressive (i.e., non-interlaced) video display from aninterlaced video signal, the video signal must be de-interlaced.Conventional de-interlace methodologies can be divided into two maincategories: (1) 2D de-interlacing; and (2) 3D de-interlacing. In 2Dde-interlacing, a frame is recreated from a single field viainterpolation of the rows in that field. A common 2D de-interlacingtechnique involves duplicating each row of a single frame to providepixel values for the blank rows; i.e., each blank row in an odd fieldcould be filled with a copy of the odd row directly below that emptyrow, while each blank row in an even field could be filled with a copyof the even row directly above that empty row. The 2D de-interlacing isparticularly useful for scenes involving fast motion since even if ascene change occurs between consecutive fields, such changes would notdistort a frame formed using “pure” common-field pixel interpolation(i.e., formed using only the pixels in a single field). For additionalinformation on 2D and 3D mixing, see U.S. Pat. No. 7,142,223(application Ser. No. 10/659,772 by Zhu, issued on Nov. 28, 2006, titled“Mixed 2D and 3D de-interlacer” and assigned to Huaya Microelectronics,Ltd.), the contents of which are incorporated herein by reference.

Hence, there is a need for improved digital video decoders to minimizedistortion in a recreated image.

BRIEF SUMMARY OF THE INVENTION

Some embodiments of the present invention provide for a digital videoresampler and decoder comprising: an input buffer coupled to a source ofa video signal oversampled by at least two times; a horizontalsynchronization detector coupled to the input buffer, the horizontalsynchronization detector to detect horizontal synchronizationboundaries; a counter coupled to the horizontal synchronizationdetector, the counter to count a number of samples from between thehorizontal synchronization boundaries; a comparator coupled to thecounter, the comparator to compare the counted number of samples to areference count; a sample corrector coupled to the input buffer, whereinthe sample corrector modifies a block of samples based on a result fromthe comparator; an output buffer coupled to the sample corrector to holdthe modified block of samples; and a comb filter coupled to the outputbuffer, the comb filter to generating a first and secondthree-dimensional color values (U3D & V3D) based on the modified blockof samples.

Some embodiments of the present invention provide for a method ofdigital video decoding, the method comprising: buffering samples;detecting horizontal synchronization boundaries; counting a number ofsamples from between the horizontal synchronization boundaries;comparing the counted number of samples to a reference count; correctingsamples in the buffer based on the act of comparing the counted numberof samples to the reference count to generate a modified block ofsamples; rebuilding a horizontal synchronization signal; generating afirst three-dimensional color value (U3D) based on the modified block ofsamples; and generating a second three-dimensional color value (V3D)based on the modified block of samples.

These and other aspects, features and advantages of the invention willbe apparent from reference to the embodiments described hereinafter.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of the invention will be described, by way of example only,with reference to the drawings.

FIG. 1A is an illustration of an interlaced video signal.

FIG. 1B is a diagram of the formation of an interlaced video signal.

FIGS. 1C and 1D show an analog-to-digital converter and a clockgenerator.

FIG. 2A shows luminance separation from a CVBS signal followed bychrominance separation based on a signal without luminance.

FIG. 2B shows chrominance separation from a CVBS signal followed byluminance separation, in accordance with the present invention.

FIG. 3 illustrates a digital video decoder architecture, in accordancewith the present invention.

FIGS. 4A, 4B, 4C, 4D and 4E show inputs and outputs of logic in adigital video decoder architecture, in accordance with the presentinvention.

FIGS. 5A, 5B and 5C show elements of an analog-to-digital conversionprocess.

FIGS. 6A, 6B, 6C, 6D, 6E and 6F show various horizontal scan lines forstandard and non-standard baseband video signals.

FIGS. 7A, 7B, 7C and 7D show scan lines in relationship to digitaltiming, extraction and recovery, in accordance with the presentinvention.

FIG. 8A and 8B illustrates a process of digital timing, extraction andrecovery, in accordance with the present invention.

FIG. 9 shows synchronization logic, in accordance with the presentinvention.

FIG. 10 shows circuitry in a CVBS resampler and a timing extractor, inaccordance with the present invention.

FIG. 11 shows a horizontal synchronization slicer, in accordance withthe present.

FIGS. 12A and 12B show lines and columns of a frame, in accordance withthe present invention.

FIG. 13 relates pixels to variables, in accordance with the presentinvention.

FIG. 14 shows circuitry for generating a 2D luminance value, inaccordance with the present invention.

FIGS. 15A and 15B show lines and columns from multiple frames, inaccordance with the present invention.

FIG. 16 relates pixels to variables, in accordance with the presentinvention.

FIG. 17 shows circuitry for generating a temporal chrominance velocity,in accordance with the present invention.

FIG. 18 shows circuitry for generating temporal luminance, in accordancewith the present invention.

FIGS. 19A and 19B show circuitry for mixing spatial and temporalluminance values, in accordance with the present invention.

FIGS. 20A, 20B, 21A, 21B, 22 and 23 show circuitry for computing spatialand temporal chrominance values, in accordance with the presentinvention.

FIGS. 24 and 25 show circuitry for mixing spatial and temporal luminanceand chrominance values, in accordance with the present invention.

DETAILED DESCRIPTION OF THE INVENTION

In the following description, reference is made to the accompanyingdrawings which illustrate several embodiments of the present invention.It is understood that other embodiments may be utilized and mechanical,compositional, structural, electrical, and operational changes may bemade without departing from the spirit and scope of the presentdisclosure. The following detailed description is not to be taken in alimiting sense. Furthermore, some portions of the detailed descriptionwhich follows are presented in terms of procedures, steps, logic blocks,processing, and other symbolic representations of operations on databits that can be performed in electronic circuitry or on computermemory. A procedure, computer executed step, logic block, process, etc.,are here conceived to be a self-consistent sequence of steps orinstructions leading to a desired result. The steps are those utilizingphysical manipulations of physical quantities. These quantities can takethe form of electrical, magnetic, or radio signals capable of beingstored, transferred, combined, compared, and otherwise manipulated inelectronic circuitry or in a computer system. These signals may bereferred to at times as bits, values, elements, symbols, characters,terms, numbers, or the like. Each step may be performed by hardware,software, firmware, or combinations thereof.

This application relates to the following application each having afiling date common with the present application and each of which areincorporated by reference herein in their entirety:

U.S. patent application Ser. No. 12/352,582, filed Jan. 12, 2009,entitled “Digital Video Decoder Architecture”.

U.S. patent application Ser. No. 12/352,542, filed Jan. 12, 2009,entitled “Adaptive Gain and Offset Control in a Digital Video Decoder”.

U.S. patent application Ser. No. 12/352,607, filed Jan. 12, 2009,entitled “Multi-Directional Comb Filtering in a Digital Video Decoder”.

FIGS. 1C and 1D show a clock generator 10 and an analog-to-digitalconverter (ADC) 20. The clock generator 10 generates a synchronousclock, which is provided to the ADC 20. The ADC 20 also receives ananalog composite video baseband signal (CVBS), synchronously samples theanalog CVBS signal and outputs a digital CVBS signal. The generatedclock signal is synchronized with the CVBS signal. The clock generator10 may generate the synchronous clock based on a feed forward signal, asshown in FIG. 1C. In FIG. 1C, the clock generator 10 accepts the analogCVBS signal fed forward from the input port. Alternately, as shown inFIG. 1D, a feedback signal may supply the already digitized CVBS signalto the clock generator 10. Using a digital CVBS signal to generate thesynchronous clock allows the clock generator 10 to be implemented indigital hardware or in a microprocessor/microcontroller.

FIG. 2A shows a Y/C separator 30 and a chroma demodulator 40. In thisconfiguration, luminance separation occurs prior to chrominancedemodulation. A luminance separator (Y/C separator 30) accepts a digitalCVBS signal, which has been synchronously sampled. The Y/C separator 30extracts a luminance signal (Y) from the input digital CVBS signal. TheY/C separator 30 also extracts a chrominance signal (C) from the digitalCVBS signal. After the luminance signal (Y) has been removed, thechrominance signal (C) still contains multiple color components. Thechrominance demodulator (chroma demodulator 40), also called a colordecoder, separates color signals. For example, the chroma demodulator 40may separate a first color signal (U) from a second color signal (V) andprovide both signals as outputs.

FIG. 2B shows a chrominance demodulator (chroma demodulator 400, alsoreferred to as a color decoder, UV color decoder or color demodulator),a comb filter 600 and an optional mixer 700, in accordance with thepresent invention. Both the chroma demodulator 400 and the comb filter600 are supplied a digital CVBS signal, which in some embodiments is asynchronously sampled signal and in other embodiments is an asynchronously sampled signal. In still other embodiments, the digitalCVBS signal is a corrected CVBS signal as described in detail below. Inthe embodiment shown, the chroma demodulator 400 first separates thefirst and second chrominance signals (e.g., first color signal U′ andsecond color signal V′) and then the comb filter 600 produces aluminance signal (e.g., Y). An optional mixer 700 may be used to furtherrefine the luminance and/or chrominance signals by mixing and scalingspatial, temporal and/or filtered luminance and chrominance signals toproduce final output signals Y_(MIX), U_(MIX) and V_(MIX).

As shown in FIG. 2B, luminance separation occurs after chrominancedemodulation unlike the configuration shown in FIG. 2A. A source of CVBSis supplied to both a chrominance (chroma) demodulator 400 as well as toa comb filter 600, which may include an internal mixer 700 or a externalmixer. The chroma demodulator 400 separates raw chrominance signalcomponents (e.g., U′ & V′). The comb filter 600 accepts the raw chromasignals in addition to the corrected CVBS signal and produces a mixedluminance signal (Y_(MIX)) and two mixed chrominance values (U_(MIX) &V_(MIX)).

Unlike the previous configuration shown in FIG. 2A, in FIG. 2B the rawchrominance signals (U′ & V′) are generated first then used to generatorthe luminance signals (Y or Y_(MIX)) and possibly refined chrominancesignals (U_(MIX) & V_(MIX)).

The source of the CVBS signal may convert a gamma-corrected red, greenand blue (R′G′B′) signal to a monochrome luminance (Y) by the weightingformula, such as Y=0.299*R′+0.587*G′+0.114*B′. This source is alsoreferred to as corrected CVBS. The source may also convert the R′G′B′signal to first and second chrominance signals, which may be U & V orequivalently I & Q or other pair of orthogonal vectors orpseudo-orthogonal vectors. Once Y is determined, the color information,in the form of a first color signal and a second color signal, may bederived from the R′G′B′ signal with the following weighting formula:U=0.492*(B′−Y) and V=0.877*(R′−Y); or I=0.736*(R′−Y)−0.268*(B′−Y) andQ=0.478*(R′−Y)+0.413*(B′−Y). The source of the CVBS signal modulates thefirst and second color signals to a phasor signal. For example, the U &V (or equivalently, I & Q) values are used to modulate a subcarriersignal (e.g., f_(SC)=3,579,545 Hz≈3.58 MHz) to produce a color value C.For example, C=Q*sin(ωt+33°)+I*cos(ωt+33°), where ω=2πf_(SC).

FIG. 3 illustrates a digital video decoder architecture and FIGS. 4A,5B, 4C, 4D and 4E show inputs and outputs of logic in the digital videodecoder architecture of FIG. 3, in accordance with the presentinvention. FIG. 3 shows a converter 100, a CVBS resampler 200, a timingextractor 300, a chroma demodulator 400, a notch filter 500, a combfilter 600, and a mixer 700, each discussed in more detail below withreference to the remaining figures. These blocks (100-700) may beimplemented in a microprocessor, a microcontroller, an ASIC or a VLSIhardware and the like, as well as in combinations thereof.

The converter 100 transforms an analog CVBS signal into a digital CVBSsignal. The converter 100 received an analog composite baseband (CVBS)signal at an input port and provides a digital CVBS signal at an outputport. In some embodiments, the digital CVBS signal is synchronouslysampled as described with reference to FIGS. 1C and 1D above. In otherembodiments, the digital CVBS signal is asynchronously sampled. Theconverter 100 includes an analog-to-digital converter (ADC) and may alsocontain an automatic gain control (AGC) circuit or automatic gain andoffset control (AGOC) circuit. The converter 100 is described furtherbelow with reference to FIGS. 5A, 5B and 5C.

The CVBS resampler 200, also shown in FIG. 4A, generates a correctedCVBS signal based the digital CVBS signal received from the converter100. The CVBS resampler 200 corrects errors introduced by asynchronoussampling. Furthermore, the CVBS resampler 200 corrects errors inherentin the CVBS signal itself. For example, a CVBS signal may be elongatedin time because a video tape medium is stretched during playback. TheCVBS resampler 200 is described further below with reference to FIGS.6A, 6B, 6C, 6D, 6E, 8, 11A, 11B, 11C and 11D.

The timing extractor 300, also shown in FIG. 4B, extracts timing signalsfrom a digital CVBS signal. The timing signals may include a verticalsynchronization signal (V_(SYNC)), a horizontal synchronization signal(H_(SYNC)), a horizontal reference signal (H_(REF)), and a phase errorsignal. In some embodiments, the digital CVBS signal is a synchronousCVBS signal from the converter 100. In other embodiments, the digitalCVBS signal is an asynchronous CVBS signal from the converter 100. Instill other embodiments, the digital CVBS signal is a corrected CVBSsignal from the CVBS resampler 200. The timing extractor 300 isdescribed further below with reference to FIGS. 6A, 6B, 6C, 6D, 6E, 7,8, 9 and 10.

The chroma demodulator 400, described with reference to FIG. 2B and FIG.4C, generates color signals (e.g., U′ and V′) based on the correctedCVBS signal. The notch filter 500, also shown in FIG. 4D, filters thecorrected CVBS signal to generate a luminance signal (Y_(NOTCH)). Thenotice filter 500 attempts to remove the color spectrum (C) from theCVBS (Y+C) signal thereby primarily leaving the luminance spectrum(Y_(NOTCH)).

The comb filter 600, also shown in FIG. 4E, filters the corrected CVBSsignal to generate a luminance signal based on the color signals. Thecomb filter 600 may generate a spatial luminance (Y_(2D COMB)) and atemporal luminance (Y_(3D COMB)). Alternative embodiments of the combfilter 600 may generate a luminance value (Y_(COMB)) that pre-mixes thespactial and temporal luminance values. The comb filter 600 may alsofilter the corrected CVBS signal to generate spatial chrominance signals(e.g., U_(2D) _(—) _(COMB) and V_(2D) _(—) _(COMB)) and a temporalchrominance signals (e.g., U_(3D) _(—) _(COMB) and V_(3D) _(—) _(COMB))based on the corrected CVBS signal and the color signals (e.g., U′ andV′) from the chroma demodulator 400. Alternatively, the comb filter 600may pre-mix the spatial color signals with color demod signals (U′ & V′)from the chroma demodulator 400 to generate a set of spatial filteredcolor values (e.g., U_(2D) & V_(2D)). Alternatively, the comb filter 600may pre-mix the color signals to generate a set of comb filtered colorvalues (e.g., U_(COMB) & V_(COMB)). Alternatively, the color signals (U′& V′) may either pass through or bypass the comb filter 600. Also, delaycircuitry may be included within or external to the comb filter 600 toprovide a delayed version of the corrected CVBS signal or othergenerated luminance and/or chrominance signals. Each of thesealternative embodiments of the comb filter 600 is described furtherbelow with reference to FIGS. 12A, 12B, 13, 14, 15A, 15B, 16, 17, 18,19A-B, 20A-B, 21A-B, 22 and 23.

The mixer 700 of FIG. 3 weights and combines the luminance signals(Y_(NOTCH), Y_(2D COMB) and Y_(3D COMB), or Y_(NOTCH), Y_(COMB)) andchrominance signals (e.g., U′, U_(2D), U_(3D), V′, V_(2D) V_(3D), or U′,U_(COMB), V′, V_(COMB)) to produce a resulting luminance signal(Y_(MIX)) and set of resulting chrominance signals (e.g., U_(MIX) andV_(MIX)). The mixer 700 is described further below with reference toFIGS. 19, 24, 25, 26 and 27.

FIGS. 5A, 5B and 5C show elements of an analog-to-digital conversionprocess. FIG. 5A includes a clock generator 110 supplying anasynchronous clock to an analog-to-digital converter (ADC) 120. In thecase of asynchronous sampling, the clock signal is generated from asource that is asynchronous from timing in the CVBS signal. For example,the source may be a crystal or free running circuit such as a phaselocked loop (PLL). The analog-to-digital converter 120 uses theasynchronous clock to oversample and transform an analog CVBS signalinto a digital CVBS signal, which is thereby both asynchronously sampledand oversampled. Oversampling results when the frequency of a clock isat least twice great as the Nyquist frequency. In this case, a CVBSsignal has a bandwidth of approximately 0 to 4.2 MHz or less thanapproximately 6 MHz. Therefore, the CVBS signal would have a Nyquistfrequency of less than 12 MHz. Embodiments of the present invention usea sampling frequency of at least twice the signal's Nyquist frequency.For example, embodiments of the present invention may use a samplingfrequency between 24 MHz and 100 MHz, such as 24 MHz (2× oversampling),36 MHz (3× oversampling), 48 MHz (4× oversampling) or 60 MHz (5×oversampling). Some embodiments of the present invention may use asampling frequency that is a non-integer multiple of the Nyquistsampling frequency (e.g., 54 MHz). In some embodiments, the oversamplingfrequency is programmable.

Conventionally, sampling at a rate far above the Nyquist rate leads toinefficient use of sampling bandwidth. That is, the oversampled dataincludes extra data, which is typically unneeded. In some embodiments ofthe current invention, oversampling provides extra data used by the CVBSresampler 200. The corrected CVBS data from the CVBS resampler 200 maybe downsampled to an equivalent sampling frequency equal to or justgreater than the input signal's Nyquist rate.

FIG. 5B includes the clock generator 110 and analog-to-digital converter120 of FIG. 4A with the addition of an automatic gain and offset control(AGOC) circuit 130. The AGOC circuit 130 performs both automatic gaincontrol (AGC) and automatic offset control (AOC). Some embodimentsinclude just an AGC and not an AOC. Other embodiments include just anAOC and not an AGC, while still other embodiments include both an AGCand an AOC. The AGOC circuit 130 monitors the digital CVBS signalgenerated by the analog-to-digital converter 120 to make sure that thedynamic range of the converter 120 is more optimally utilized. Forexample, assuming the analog-to-digital converter 120 has an outputvoltage with a peak-to-peak voltage (V_(PP)) of 1 Volt. If an incominganalog CVBS signal shows a peak-to-peak voltage of 0.1 Volts (i.e., 10%of dynamic range of converter 120), the AGOC 130 may provide a controlsignal to the converter 120 to increase its amplitude by a factor often. Similarly if the incoming CVBS signal shows a DC offset that is toohigh or too low, the AGOC 130 may provide a second control signal to theconverter 120 to adjust the offset seen at the output of the converter120.

FIG. 5C shows a crystal (xtal 112) and a phase locked loop (PLL 114), inaccordance with the present invention. In the embodiment shown, thecrystal 112 provides a 12 MHz (e.g., with a stability of ±100 PPM) tothe PLL 114. The PLL 114 may be programmable to produce variousfrequencies or may be designed to produce a single frequency (e.g., 54MHz as shown). A clock produced in isolation from the input signal willbe asynchronous from the input signal and thus termed an asynchronousclock.

FIGS. 6A, 6B, 6C, 6D and 6E show various horizontal scan lines forstandard and non-standard baseband video signals.

In FIG. 6A, a standard-dimensioned horizontal scan line from a CVBSsignal is shown. The standard horizontal scan line signal has apeak-to-peak voltage (Δ₁=V_(PP)=V_(MAX)−V_(MIN)) of 1 Volt, a sync pulse(V_(SYNC)=V_(REF)−V_(MIN)) of 300 millivolts (mV) below a blanking level(V_(REF)), and a maximum envelope voltage (V_(ENV)=V_(MAX)−V_(REF)) of700 mV above the blanking level (V_(REF)). Using a 10-bit ADC, theminimum is shown at symbol 2 and may be identified as a digital code 0from the ADC 120. The blanking level (V_(REF)) is shown at symbol 4 andmay be identified as a digital code 307. The maximum voltage (V_(MAX))is shown at symbol 6 and may be identified as a digital code 1024. Thedifference between the blanking level (code 307) and the maximum (code1023) may be referred to as Δ₂ or the envelope voltage (V_(ENV)). Thedifference between the blanking level (code 307) and the minimum (code0) may be referred to as Δ₃ or the sync level (V_(SYNC)). A standardsignal will also have a ratio of Δ₂:Δ₃ of 7:3 from the desired valuesV_(ENV) _(—) _(DESIRED)=700 mV to V_(SYNC) _(—) _(DESIRED)=300 mV.

The horizontal synchronization pulse (sync pulse), at symbol 1, startsat the blanking level V_(REF), falls to the minimum voltage V_(SYNC) for4.7 microseconds (μs), and then returns to the blanking level V_(REF). Acolor burst, at symbol 3, follows the sync pulse. The color burst is asubcarrier burst with a duration of approximately 5.6 μs (PAL) or 5.3 μs(NTSC). The luminance and chrominance data is encoded in an envelope, atsymbol 5. The luminance and chrominance envelope varies in amplitude(V_(ENV)) depending on the encoded luminance and chrominance data.

FIG. 6B shows two lines of encoded luminance and chrominance data. Inthe first envelope, a dark scene, at symbol 7, is shown having anenvelope magnitude of V_(ENV)=300 mV, which happens to be the samemagnitude of the sync pulse (V_(SYNC)). A subsequent line has anenvelope, at symbol 8, containing luminance and chrominance datarepresenting a bright scene. This envelope has an amplitude ofV_(ENV)=700 mV above the blanking level. The figure illustrates that theluminance and chrominance envelopes vary in amplitude. As a consequence,a frame containing a sequence of lines with dark scenes may appear ashaving non-standard ratio of Δ₂:Δ₃, when the lines actually havestandard signal encoding. Over time, the envelop voltage (V_(ENV)) maybe monitored to find its maximum, which will occur during bright scenes.

FIG. 6C shows a first non-standard signal. The first non-standard signalhas a standard peak-to-peak voltage (V_(PP)=1 V) however the ratio ofΔ₂:Δ₃ of 8:2 is beyond the possible ratio of 7:3 maximum of a standardsignal. Determining that this signal is a non-standard signal may occurafter observation of a single line of data. That is, a standard signalmay never have an envelope that has a maximum amplitude that is morethan 7/3 of the amplitude of the sync pulse. To transform this firstnon-standard signal into a standard signal requires forcing the ratioback to 7:3 by attenuating the signal during the luminance andchrominance envelope from a maximum of V_(ENV)=800 mV to 700 mV and alsoamplifying the sync pulse from V_(SYNC)=200 mV to 300 mV. In this case,the resulting transformed signal has a blanking level V_(REF) that hasbeen shifted up by 100 mV but keeps it peak-to-peak voltage of 1 V.

FIG. 6D shows a second non-standard signal. The second non-standardsignal also has a standard peak-to-peak voltage (V_(PP)=1 V), however,the ratio of Δ₂:Δ₃ of 6:4 is below the standard ratio of 7:3. After asingle observation, one cannot determine whether the line represent adark scene or is from a non-standard CVBS signal. After observingseveral lines of data though, one expects to receive a variety dark andbright scenes. Therefore, if the maximum ratio ever received is 6:4,then the received signal represents a non-standard CVBS signal. Totransform this second type of non-standard signal into a standard signalrequires a similar procedure of forcing the ratio back to 7:3. Thesignal may be amplified during the luminance and chrominance envelopefrom its maximum of 600 mV to 700 mV and attenuated during the syncpulse from 400 mV to 300 mV. In this case, the resulting transformedsignal has a blanking level the has been shifted down by 100 mV but alsokeeps it peak-to-peak voltage of 1 V.

FIG. 6E shows a standard but weak signal. The signal has a below anon-standard peak-to-peak voltage (V_(PP)=0.1 V) but fortunately theratio of Δ₂:Δ₃ of 7:3. Simple 10× amplification of the signal withappropriate offset control results in an adjusted signal having below astandard peak-to-peak voltage (V_(PP)=1 V) with a maintained ratio ofΔ₂:Δ₃ of 7:3.

In some embodiments, the AGOC circuit 130 of FIG. 5B using a differentset of amplitude and offset controls for each of the sync pulse periodand the envelope period. In other embodiments, the AGOC circuit 130 ofFIG. 5B using a common set of amplitude and offset controls for periodof the CVBS signal including both the sync pulse period and the envelopeperiod. In these embodiments, the AGOC circuit 130 selects the gain andoffset control values based on the detected V_(REF) and V_(ENV) levelswithout regard to the V_(SYNC) level. If the Δ₂:Δ₃ ratio is less than7:3, as in the example of FIG. 6D, the bottom of the sync pulse will beclipped to thus maintaining a V_(PP) less than or equal to 1 V.

The AGOC circuit 130 of FIG. 5B may monitor the output CVBS signal fromthe ADC 120 to measure signal parameters. The signal parameters may beused by the AGOC circuit 130 to adjust the amplitude and offset controlsused by the ADC 120. The signal parameters may include: (1) a measure ofthe blanking level, V_(REF); and (2) a measure of the maximum envelopvoltage, V_(ENV), with reference to the blanking level. These signalparameter may be measured over a period of time after receiving severallines of data. The AGOC circuit 130 uses these signal parameters todetermine an offset and a gain to apply to the incoming CVBS signal,thereby resulting in a standard CVBS signal at the output port of theADC 120.

FIG. 6F shows a process to maintain and update gain and offset controls.At step 141, a gain parameter and an offset parameters, used by the ADC120, are initialized to default levels. At step 142, the ADC 120converts an analog CVBS signal into a digital CVBS signal based on thegain and offset parameters. At step 143, the AGOC 130 monitors thedigital CVBS signal to determine a blanking voltage V_(REF). With thedigital signal, the blanking voltage will be represented by a code(e.g., code 307 from a 10-bit ADC). At step 144, the AGOC 130 determinesan envelop voltage, which is the difference between the maximum signallevel (V_(MAX)) and the blanking signal level (V_(REF)). Again, themaximum and blanking signal levels will be represented by other codes.The maximum signal level may be determined after several measurements ofthe envelope to increase the probability that at least one bright sceneis encountered. The blanking signal level may be determined by examininga single line, for example, by determining at what level the signalresides when not within a sync pulse, a color burst or luminance andchrominance data envelope.

At step 145, the AGOC 130 determines a new gain for the ADC 120. Thegain may be adjusted quickly to a determined gain or may be adjustedgradually, using a low pass filter of a sequence of determined gainvalues. The next gain (G_(NEXT)) may be set to the previous gain(G_(PREV)) scaled by the ratio of the desired envelope amplitude(V_(ENV) _(—) _(DESIRED)) to the actual envelop amplitude (V_(ENV)), orG_(NEXT)=G_(PREV)*(V_(ENV) _(—) _(DESIRED)/V_(ENV)).

At step 146, the AGOC 130 determines a new offset to drive the blankingvoltage (V_(REF)) to a desired blanking voltage (V_(REF) _(—)_(DESIRED)). Again, the offset may be adjusted directly to a determinedoffset or may be adjusted gradually, using a low pass filter of asequence of determined offset values. The next offset (OFFSET_(NEXT))may be set to the previous offset (OFFSET_(PREV)) adjusted by the errorbetween the desired blanking amplitude (V_(REF) _(—) _(DESIRED)) to themeasured blanking amplitude (V_(REF)), orOFFSET_(NEXT)=OFFSET_(PREV)+(V_(REF) _(—) _(DESIRED)−V_(REF)).

For example, assume the previous gain (G_(PREV)) was unity, the previousoffset (OFFSET_(PREV)) was 0.0, the desired envelope amplitude (V_(ENV)_(—) _(DESIRED)) is =0.7 V (relative code 716=1023−307) and the desiredblanking amplitude (V_(REF) _(—) _(DESIRED)) is 0.3 V (code 307).Further assuming that the non-standard CVBS signal of FIG. 6C isreceived, with V_(PP)=1 V, V_(SYNC)=0.2 V, V_(REF)=0.2 V, andV_(ENV)=0.8 V. The gain indicator may be computed as: G_(NEXT)=G_(PREV)*(V_(ENV) _(—) _(DESIRED)/V_(ENV))=1*(0.7/0.8)=0.875 (an attenuation).The next gain may be set directly to the determined gain 0.875 or may beset to a value that approaches this determined gain. For example, thenext gain may be an average of the previous gain (unity) and thedetermined gain (0.875), which would be (1+0.875)/2=0.9375 (a slightattenuation but not the full determined attenuation thereby reducingoscillations). Alternatively, another LPF could have been used to filtera sequence of determined gains to compute a gain that will be providedto the AGC 120. Similarly, the offset indicator may be computed as:OFFSET_(NEXT)=OFFSET_(PREV)+(V_(REF) _(—)_(DESIRED)−V_(RE)F)=0+(0.3−0.2)=0.1 (a step up). Again, this offset maybe used directly or may be filtered. For example, an average between theprevious offset and the determined value (0.1+0.0)/2=0.05 may be usedfor the next offset.

As another example, assume that the non-standard CVBS signal of FIG. 6Dis received, with V_(PP)=1 V, V_(SYNC)=0.4 V, V_(REF)=0.4 V, andV_(ENV)=0.6 V. The gain indicator may be computed as:G_(NEXT)=G_(PREV)*(V_(ENV) _(—) _(DESIRED)/V_(ENV))=1*(0.7/0.6)=1.167(an amplification). Similarly, the offset indicator may be computed as:OFFSET_(NEXT)=OFFSET_(PREV)+(V_(REF) _(—)_(DESIRED)−V_(REF))=0+(0.3−0.4)=−0.1 (a step down).

As a further example, assume that the standard but weak CVBS signal ofFIG. 6E is received, with V_(PP)=0.1 V, V_(SYNC)=0.03 V, V_(REF)=0.03 V,and V_(ENV)=0.07 V. The gain indicator may be computed as:G_(NEXT)=G_(PREV)*(V_(ENV) _(—) _(DESIRED)/V_(ENV))=1*(0.7/0.07)=10 a(a10× amplification). Similarly, the offset indicator may be computed as:OFFSET_(NEXT)=OFFSET_(PREV)+(V_(REF) _(—)_(DESIRED)−V_(REF))=0+(0.3−0.03)=0.297 (a step up).

In some embodiments, a gain indicator and an offset indicator are set todrive the voltage between the blanking level and the maximum level ofthe signals to a first standard voltage (e.g., 0.7 V) and the blankinglevel to a second standard voltage (e.g., 0.3 V).

For example, the AGOC 130 measures a blanking level and a maximum levelof a CVBS signal. The AGOC 130 compares the blanking level to a firsttarget range for acceptable voltages of signal's blanking level (e.g.,within a predetermined percentage of 0.3 V, within a range centered at areference voltage of 0.3 volts, or within a range centered at areference voltage of 30% of a dynamic range of the ADC).

The AGOC 130 also compares the maximum level to a second target rangefor acceptable maximum voltages (e.g., within a predetermined percentageof a maximum voltage of 1.0 V). The blanking level or the maximum levelis outside the acceptable ranges, the AGOC 130 adjusts a gain indicator.That is, if the blanking level is outside a first target range or themaximum level is outside a second target range then the AGOC 130 eitherincreases or decreases the gain indicator feed to the ADC 120. Forexample, if the blanking level is below the first target range and themaximum level is below the second target range the AGOC 130 increasesthe gain indicator. If the blanking level is above the first targetrange or if the maximum level is above the second target range, the AGOC130 decreases the gain indicator. In this way, the AGOC 130 adjusts theblanking and maximum levels independent of the signal's minimum level. Aresulting signal has a blanking level within the first target range anda maximum level within the second target range. Since the maximum leveland the blanking level are driven to particular voltages, the minimumlevel will be somewhere below the blanking level but not driven to aparticular voltage.

After the AGOC 130 determines the gain and offset values, it sendsupdated gain and offset indicators to the ADC 120. These values may beabsolute values as described above or may be step controls to indicatethat a value should be incrementally increased or decreased as indictedby the step controls. In either case, the process of determining a gainand an offset is iterative, as shown by step 146 returning to step 142.In a steady state condition, the gain and offset indicators settle tofixed values and the maximum and blanking amplitudes of the digitizedsignal approach those of a standard signal.

In some embodiments, a gain is adjusted based on the maximum level andthe blanking level as follows. If either a maximum level or a blankinglevel is too high, a subsequent gain is decreased. That is, if themaximum level is above a first target range or the blanking level isabove a second target range, then the gain is decreased. If both themaximum level and blanking level are too low, a subsequent gain isincrease. That is, if the maximum level is below the first target rangeand the blanking level is below the second target range, then the gainis increase. In all other cases, the gain is kept constant.

In some embodiments, a gain is adjusted based on a difference betweenthe maximum level and the blanking level and based on the blanking levelas follows. The maximum level and blanking level are determined. Next adifference is computed between the maximum level and blanking levels. Ifeither the difference or the blanking level is too large, a subsequentgain is decreased. That is, if the difference is above a first targetrange or the blanking level is above a second target range, then thegain is decreased. If both the difference and blanking level are toolow, a subsequent gain is increase. That is, if the difference is belowthe first target range and the blanking level is below the second targetrange, then the gain is increase. In all other cases, the gain is keptconstant.

In some embodiments, an offset is adjusted based on a blanking level. Ifthe blanking level too low, then the offset is increased. That is, ifthe blanking level is below a blanking target range (e.g., a valuewithin an error of 0.3 V), then the offset is increased. If the blankinglevel too high, then the offset is decreased. That is, if the blankinglevel is above the blanking target range, then the offset is decreased.

In automatic gain control (AGC) embodiments, only a gain is adjusted andan offset control is not available. In automatic offset control (AOC)embodiments, only an offset is adjusted and gain control is notavailable. In automatic gain and offset control (AGOC) embodiments, botha gain and an offset are available for adjusting.

Changes to a gain or an offset may take priority over the other. In someembodiments, an offset is not adjusted if a gain needs to be adjusted.In these embodiments, adjustment of the gain, as described above, takespriority over adjustment of the offset. In other embodiments, a gain isnot adjusted if an offset needs to be adjusted. In these embodiments,adjustment of the offset, as described above, takes priority overadjustment of the gain. In still other embodiments, the both the gainand the offset may be adjusted simultaneously.

To correct amplitude domain errors in the input signal, the AGOC 100determines the gain and offset indicators as described above. To correcttime domain errors (and phase errors) in the input signal, the CVBSresampler 200 may resample in the input signal as described in furtherdetail below with reference to FIGS. 7A-11. The CVBS resampler 200extracts horizontal and vertical timing information, used for systemsynchronization, from the digital CVBS signal. The CVBS resampler 200uses the horizontal sync pulse from line to line to determine whether aline of data is of standard length or non-standard length. A line ofdata that is of non-standard length has either too few samples or toomany samples. A non-standard line may occur from reading line data froma video tape medium. For example, the tape may be stretched, therebyresulting in too many samples. Alternatively, the tape play may bepaying back data too quickly, thus resulting in too few samples. Anon-standard line may also occur if segments of data where not receivedby the decoder.

FIGS. 7A, 7B, 7C and 7D show scan lines in relationship to digitaltiming, extraction and recovery, in accordance with the presentinvention. The CVBS resampler 200 may correct errors in sample timingwithin the raw digitized CVBS signal by a process of interpolation,expansion and/or compression.

FIG. 7A illustrates samples from a horizontal scan line of a CVBSsignal. On the left, the figure shows a full line including a first syncpulse delineating the beginning of the line and a second sync pulsedelineating the end of the line. On the right, the figure shows foursample points made by the converter 100. A standard signal length has aduration between sync pulses of the inverse of the horizontal linefrequency. A line of CVBS data having a duration between sync pulses ofapproximately 63.6 to 64 μs (corresponding to 15.725 kHz for NTSC and15.625 kHz for PAL) within a threshold may be considered to be a CVBSsignal of standard period.

FIG. 7B illustrates samples from a horizontal scan line that is tooshort thus has too few samples. On the left, the figure shows a measuredduration between sync pulses that shows a short duration line signal,which needs to be expanded to have additional samples. On the right, thefigure shows that an original set of four samples has been replaced withfive samples. Samples may be added by resampling the digital signal atan effective higher sampling rate needed to insert the desired number ofsamples. This resampling may be limited to the luminance and chrominanceportion of the signal. Thus, the new resampled signal will have a propernumber of samples within the envelope segment of the line signal.

FIG. 7C illustrates samples from a horizontal scan line of a CVBS signalthat is too long thus has too many samples. On the left, the figureshows a measured duration between sync pulses that shows a long durationline signal, which needs to be compressed to have reduce the totalnumber of samples. On the right, the figure shows that an original setof four samples has been replaced with three samples. Samples may beremoved by resampling the digital signal at an effective lower samplingrate needed to reduce the sample count to the desired number of samples.Again, the resampling may be limited to the luminance and chrominanceportion of the signal.

As described above, a set of samples representing the luminance andchrominance envelope may be resampled to compress or expand the sampleddata to form an envelope of standard length. When the data is ofstandard length, the luminance and chrominance data may be extractedfrom the envelope as described below. In some embodiments, the phase ofthe samples is also changed. In some embodiments, the phase is checkedand changed after the signal has been expanded or compressed. In otherembodiments, the phase is checked only if it was unnecessary to expandor compress the signal. In other embodiments, the phase is checked onlyif sampled data has a number of samples within a threshold.

FIG. 7D illustrates samples from a horizontal scan line of a CVBS signalhaving a standard length. The samples are out of phase an ideal samplingtime. The CVBS resampler 200 resamples the digital CVBS signal such thatthe new samples have a correct delay with respect to a reference point,for example, from the horizontal sync pulse. The CVBS resampler 200 mayuse an interpretation process to generate the new samples. For example,if the correct timing is 30% past one sample and 70% before the nextsample, a simple line fitting algorithm may be used. Alternatively, asecond or higher order curve fitting algorithm may be used. On the rightof the figure, a line fitting algorithm has resulted in new samplesformed from interpolation between pairs of samples.

In some embodiments, resampling to expand, compress and/or interpolatemay include decimation to reduce the oversampled signal data to a signalsampled between the Nyquist rate and less than two times the Nyquistrate. Before a line of data is interpolation, expansion and/orcompression, an accurate measure of the period between pairs of syncpulses is needed. This horizontal line timing may be determined from thedigitized CVBS signal.

FIGS. 8A and 8B illustrate a process of digital timing, extraction andrecovery, in accordance with the present invention. FIG. 8A shows aprocess in a digital video resampler and decoder leading up tointerpolation, expansion and compression. At step 280, a horizontal lineof sample is buffered into memory of an input buffer. This input bufferis coupled to a source of a video signal oversampled by at least twotimes and may contain more samples than contained in a single horizontalline. At step 282, the horizontal synchronization (H_(SYNC)) aredetected by a horizontal synchronization detector coupled to the inputbuffer. The horizontal synchronization detector detect horizontalsynchronization boundaries. A first boundary may have been detected withprocessing of an immediately preceding horizontal scan line. Similarly,a second boundary detected in the current scan line may be used whenprocessing the immediately following horizontal scan line. At step 284,a counter, coupled to the horizontal synchronization detector, counts anumber of samples from between the detected horizontal synchronizationboundaries. This counter may count samples after both the first andsecond H_(SYNC) boundaries have been established. Alternatively, thecounter may count samples either as they arrive into the buffer orduring detection of the second horizontal synchronization boundary. Atstep 286, a comparator, coupled to the counter, compares the countednumber of samples to a reference count. The reference count may be thedesired number of samples (e.g., 3432 with 4× oversampling) or may be arange of acceptable samples (e.g., =3424 to 3440=3432±8).

At step 288, a sample corrector, such as CVBS resampler 200 that iscoupled to the input buffer, modifies a block of samples based on acomparison result from the comparator. For example, if the comparatorshows too many samples exist between the first and second H_(SYNC)signals, then the sample corrector compresses the number of samples.Similarly, if too few samples exists, then the sample corrector expandsthe number of samples. In either case or when the number of samples iswithin a desirable range, the samples may be interpolated to correct thetiming resulting from asynchronous sampling or signal mistiming. Steps286 and 288 are described further below with reference to FIG. 8B.

At step 290, a sampling frequency of the PLL 114 may be adjusted tobetter track the desired clock frequency and/or phase. At step 292, anH_(SYNC) signal may be rebuilt. At step 294, a comb filter 600 may beused to generate a temporal luminance signal (e.g., Y_(3D) _(—)_(COMB)). The comb filter 600 may also be used to generate a first andsecond temporal chrominance signals (e.g., U_(3D) _(—) _(COMB) & Y_(3D)_(—) _(COMB)). Furthermore, the comb filter 600 may be used to generatespatial luminance and chrominance signals (e.g., Y_(2D) _(—) _(COMB),U_(2D) _(—) _(COMB) & U_(2D) _(—) _(COMB)).

FIG. 8B shows two alternative implementations of steps 286 and 288. Atstep 286, a comparator, coupled to the counter, compares the countednumber of samples to a reference count. The reference count may be thedesired number of samples (e.g., 3432 with 4× oversampling) or may be arange of acceptable samples (e.g., =3424 to 3440=3432±8, where ±8 is thethreshold). Alternatively, the threshold may be set to zero (0) oranother value representing a percentage error (e.g., a threshold of ±1%at 4× oversampling represents ±34 samples; a threshold of ±0.5%represents ±17 samples; and a threshold of ±0.1% represents ±3 samples).If a desired number of samples within the threshold exists in thecurrent line of digital CVBS data, then follow skips step 288 andcontinues to step 288.8. If the counted number of samples is outside thethreshold, then flow continues to step 288, which includes steps 288.2,288.4 and 288.6. At step 288.2, a determination is made as two whether acounted number of samples between two reference points (e.g., betweenthe first and second horizontal synchronization pulses) above (too manysamples) or below (too few samples) the threshold count of an ideal orexpected number of samples. If too few samples exist, then processingcontinues at step 288.4. At step 288.4, the current line of samples isexpanded such that a desired number of samples exists. For example, thecurrent samples may be resampled to increase the total number of samplesin a line. Alternatively, if too many samples exist, then processingcontinues at step 288.6. At step 288.6, the current line of samples iscompressed such that a desired number of samples exists. For example,the current samples may be resampled to decrease the total number ofsamples in a line. In a first embodiment, steps 288.4 and 288.6 exit toexecute step 288.8. In a second embodiment, steps 288.4 and 288.6 exitand bypass step 288.8. Flow then continues to step 290 in FIG. 8A. As aresult of processing steps 288.4, 288.6 and 288.8, the sample correctorfills an output buffer with a modified block of samples. If noadjustment to the samples was made, the output buffer may linked to theinput samples.

FIG. 9 shows synchronization logic, in accordance with the presentinvention. The synchronization logic, such as the timing extractor 300in FIG. 3, includes a horizontal synchronization slicer (H_(SYNC) slicer310) and a vertical synchronization slicer (V_(SYNC) slicer 360). BothH_(SYNC) slicer 310 and V_(SYNC) slicer 360 accept an uncorrecteddigital CVBS signal. That is, raw samples are supplied to slicers 310and 360 rather than samples corrected by the CVBS resampler 200. TheH_(SYNC) slicer 310 analyzes the input samples to detect horizontalsynchronization pulse (H_(SYNC)), horizontal reference pulses (H_(REF))and phase error. In turn, the H_(SYNC) slicer 310 provides thesegenerated signals as reference signals to circuitry internal to the CVBSresampler 200, comb filter 600 and/or mixer 700. Similarly, the V_(SYNC)slicer 360 analyzes the input samples to detect and generate a verticalsynchronization pulses (V_(SYNC)).

FIG. 10 shows circuitry in a CVBS resampler 200 and a timing extractor300, in accordance with the present invention. In the embodiment shown,the CVBS resampler 200 includes an input buffer 202 coupled to accept aninput stream of sampled CVBS data. The input stream of sampled CVBS datamay be asynchronously sampled. In other embodiments, the input stream ofsampled CVBS data contains synchronously sampled data.

The CVBS resampler 200 also includes a corrector 204, which includes aninterpolator 205 and a resampler 206. The interpolator 205 may operateto either expand or compress the number of samples (respectively inFIGS. 7B and 7C and steps 288.4 and 288.6 of FIG. 8B) and may use afirst-degree or second-degree curve fitting algorithm or the like. Theresampler 206 may operate to adjust the phase of the samples (in FIG. 7Dand step 288.8 of FIG. 8B). The input buffer 202 supplies its data on afirst data bus to the corrector 204. In some embodiments, data from theinput buffer 202 is either interpolated by the interpolator 205 or phaseadjusted by the resampler 206. Logic may make a decision on whether tointerpolate or adjust phase based on the extent of the temporal error inthe input signal. For example, if the input data contains a total numberof samples within a threshold count of a standard signal, then the datamay go through a phase adjusting process. If the input data contains toomany or too few samples, then the data may go through an interpolationprocess. In other embodiments, data from the input buffer 202 is bothinterpolated by the interpolator 205 and phase adjusted by the resampler206. For example, a line of input data may be first undergo aninterpolation process then undergo a resampling process.

Additionally, the CVBS resampler 200 includes an output buffer 208 tohold the resulting output data from the interpolator 205 and theresampler 206. A second data bus may be used to couple the output datafrom the corrector 204 to the input of the output buffer 208.Functionally, the first and second data buses may be the same physicalbus but used at different times. Similarly, the input buffer 202 and theoutput buffer 208 may share the same memory hardware. The output of theoutput buffer 208 provides a corrected CVBS set of samples to the combfilter 600 and other functional blocks (as shown in FIG. 3).

The timing extractor 300 includes a PPL controller 370 coupled to thePLL 114 of FIG. 5C. The timing extractor 300 also includes a horizontalsynchronization slicer (H_(SYNC) slicer 310) including an H_(SYNC)filter 312, an H_(SYNC) detector 320 and an H_(SYNC) generator/smoother322. The H_(SYNC) filter 312, as shown in FIG. 11, includes aconvolution circuit 313 that accepts and convolves the digital CVBSsignal with a pulse model signal 314. The H_(SYNC) filter 312 alsoincludes an integration circuit 315 that integrates the digital CVBSsignal. The output signals from the convolution circuit 313 and theintegration circuit 315 are combined by a combiner 316, which provides aresulting filtered CVBS signal used by the H_(SYNC) detector 320.

The H_(SYNC) detector 320 detects a horizontal synchronization pulse byanalyzing the output of the H_(SYNC) filter 312. For example, when thefiltered data falls below a threshold for a number of samples, theH_(SYNC) dectector 320 may select the first, middle or last sample torepresent the H_(SYNC) transition. The resulting transition is suppliedto the H_(SYNC) generator/smoother 322 as a raw H_(SYNC) signal.

The H_(SYNC) generator/smoother 322 uses the raw H_(SYNC) signal as aninput to a PLL thereby generating a steady H_(SYNC) signal. The H_(SYNC)signal broadcasted is a periodic signal except during a verticalsynchronization period. During the vertical synchronization period, noH_(SYNC) signal is transmitted and therefore the raw H_(SYNC) signalwill not show a transition. A hold signal generated from the V_(SYNC)slicer 320 described below switches the H_(SYNC) generator/smoother 322from a state driven by the raw H_(SYNC) signal to a free running stateholding the current PLL phase and frequency of the provided H_(SYNC)signal. For a standard PAL signal, this generated H_(SYNC) signal isdesigned to operate at or 15.625 kHz. For a standard NTSC signal, thisgenerated H_(SYNC) signal is designed to operate at 15.725 kHz.

The timing extractor 300 also includes a vertical synchronization slicer(V_(SYNC) slicer 360) including a V_(SYNC) filter 362, a V_(SYNC)detector 370 and a V_(SYNC) generator/smoother 372. The V_(SYNC) filter362 filters the incoming digital CVBS signal to provide a filtered CVBSsignal to the V_(SYNC) detector 370. The V_(SYNC) detector 370 detectstransitions corresponding to a vertical synchronization signal andprovides this detected transition as a raw V_(SYNC) signal to theV_(SYNC) generator/smoother 372. The generated V_(SYNC) signal may beused as a control signal for the circuitry shown in FIG. 3. For astandard PAL signal, this generated V_(SYNC) signal is designed tooperate at 50 Hz. For a standard NTSC signal, this generated V_(SYNC)signal is designed to operate at 60 Hz.

FIGS. 12A and 12B show lines and columns of a frame, in accordance withthe present invention. FIG. 12A shows an array of pixel positionsassociated with a current field 602 from a CVBS signal within an NTSCbroadcasting system. The array of pixels shown include columns m−2, m−1,m, m+1 and m+2, where column m is a column of current interest. Thearray of pixels shown also includes several lines of pixels including aprevious line 611, a current line 612 and a next line 613. At theintersections of the column of interest (column m) and the lines ofinterest (611, 612 and 613) are pixel positions of interest. The pixelposition in the current column m and in the previous line 611 is labeledpixel p as the previous pixel. The pixel position in the current columnm and in the current line 612 is labeled pixel c as the center pixel.The pixel position in the current column m and in the next line 613 islabeled pixel n as the next pixel.

FIG. 12B shows a similar array of pixel positions associated with acurrent field 602 from a CVBS signal within a PAL broadcasting system.The array of pixels shown includes columns m−2, m−1, m, m+1 and m+2,where column m is a column of current interest. The array of pixelsshown also includes several lines of pixels including a previous line611, a current line 612 and a next line 613. At the intersections of thecolumn of interest (column m) and the lines of interest (611, 612 and613) are pixel positions of interest. The pixel positions in the currentcolumn m and in lines 611, 612 and 613 are labeled pixel p, c and nrespectively. Between pixels p and c and between pixels c and d areneighboring pixel u (“up”) and neighboring pixel d (“down”),respectively.

FIG. 13 relates data associated with pixel positions p, c and n tovariables, in accordance with the present invention. Variable cvbs0 621represents corrected CVBS data associated with the current field 602,the previous line 611 and the current column m. Variable cvbs1 622represents corrected CVBS data associated with the current field 602,the current line 612 and the current column m. Variable cvbs2 623represents corrected CVBS data associated with the current field 602,the next line 613 and the current column m. Processed CVBS dataassociated pixel positions p, c and n are used for computing mixingspatial coefficients, luminance and chrominance values as described infurther detail below.

FIG. 14 shows circuitry for generating a 2D luminance value, inaccordance with the present invention. The 2D luminance value (Y_(2D)_(—) _(COMB) 629) is based on the variables cvbs0 621, cvbs1 622 andcvbs2 623 as well as a mixing coefficient (K_(—)2dcomb 620). In theembodiment shown, average luminance values are computed then mixedtogether. The first luminance value 625 is computed using a firstaveraging unit 624 including a summer to add cvbs0 621 and cvbs1 622values followed by a divider to provide the resultY_(—)2dcomb_up=(cvbs0+cvbs1)/2. The second luminance value 627 iscomputed using a second averaging unit 626 including a summer to addcvbs1 622 and cvbs2 623 values followed by a divider to provide theresult Y_(—)2dcomb_down=(cvbs1+cvbs2)/2. Base on a mixing value(K_(—)2dcomb 620), which is inclusively between zero and one, the firstand second luminance values are scaled then summed with mixer 628. Theresulting 2D luminance value (Y_(2D) _(—) _(COMB) 629) may be written asY_(2D) _(—)_(COMB)=K_(—)2dcomb_up*Y_(—)2dcomb_up+(1−K_(—)2dcomb_up)*Y_(—)2dcomb_down.

The mixing coefficient K_(—)2dcomb 620 weights the line above thecurrent line with the line below the current line. If K_(—)2dcomb 620 isgreater than 0.5, more weight is given to the previous line than thenext line. Similarly, if K_(—)2dcomb 620 is less than 0.5, more weightis given to the next line than the previous line. Each of the mixingcoefficients described below similarly provide a weighting to data froma previous and next line or from a previous and next frame.

FIGS. 15A and 15B show lines and columns from multiple frames, inaccordance with the present invention. FIG. 15A shows three arrays ofpixel positions associated with a previous field 601, a current field602 and a next field 603 from a CVBS signal within an NTSC broadcastingsystem. The first array of pixel positions includes pixels in a column mof a previous field 601, including a previous pixel p from a previousline, a current pixel c from a current line and a next pixel n from anext line. Similarly, a second array of pixel positions includes pixelsin a column m of a current field 602, including a previous pixel p froma previous line, a current pixel c from a current line and a next pixeln from a next line. The third array of pixel positions includes pixelsin a column m of a next field 603, including a previous pixel p from aprevious line, a current pixel c from a current line and a next pixel nfrom a next line. When a current field 602 is an odd field, then theprevious field 601 and the next field 603 are also odd fields. When acurrent field 602 is an even field, then the previous field 601 and thenext field 603 are also even fields. FIG. 15B shows that a CVBS signalwithin a PAL broadcasting system have frames of interest separated by anintermediate field.

FIG. 16 relates pixels to variables, in accordance with the presentinvention. Variable cvbs_p 631 represents corrected CVBS data associatedwith the previous field 601, the current line 612 and the current columnm. Variable cvbs_c 632 represents corrected CVBS data associated withthe current field 602, the current line 612 and the current column m.Variable cvbs_n 633 represents corrected CVBS data associated with thenext field 603, the current line 612 and the current column m. TheseCVBS data are used for computing temporal mixing coefficients, luminanceand chrominance values as described in further detail below.

FIG. 17 shows circuitry for generating a temporal chrominance velocity640, in accordance with the present invention. The temporal chrominancevelocity 640 is a function of data (cvbs_p 631, cvbs_c 632 and cvbs_n633) from the three pixel positions described above with reference toFIG. 16. Respective differences are computed by units 634 and 636.Specifically, a first difference C_(—)3dcomb_p is computed by summer 634as C_(—)3dcomb_p=cvbs_p−cvbs_c. A second difference C_(—)3dcomb_n iscomputed by summer 636 as C_(—)3dcomb_n=cvbs_c−cvbs_n. These differencesare summed at 638 then a ratio between the second difference and thesummed results are taken at 639 resulting in the temporal chrominancevelocity 640, which may be written as:

${{K\_}\; 3\;{dcomb}} = {\frac{{C\_}3\;{dcomb\_ n}}{{{C\_}3\;{dcomb\_ p}} + {{C\_}3\;{dcomb\_ n}}}.}$

In other embodiments, the temporal chrominance velocity 640 may besimilarly written as:

${{K\_}\; 3\;{dcomb}} = {\frac{{C\_}3\;{dcomb\_ p}}{{{C\_}3\;{dcomb\_ p}} + {{C\_}3\;{dcomb\_ n}}}.}$

The temporal chrominance velocity 640 may be used to mix computedluminance values as described below if additional detail.

FIG. 18 shows circuitry for generating a temporal luminance value(Y_(3D) _(—) _(COMB) 646), in accordance with the present invention. Thetemporal luminance Y_(3D) _(—) _(COMB) 646 is determined based on cvbs_p631, cvbs_c 632 and cvbs_n 633. A first luminance value is computedusing summer 641 as:

${{Y\_}3\;{dcomb\_ p}} = {\frac{{cvbs\_ c} + {cvbs\_ p}}{2}.}$

A second luminance value is computed using summer 643 as:

${{Y\_}3\;{dcomb\_ n}} = {\frac{{cvbs\_ c} + {cvbs\_ n}}{2}.}$

A mixer 645 scales the first and second luminance values based on thetemporal chrominance velocity (K_(—)3dcomb 640) and then sums the scaledvalues. The resulting temporal luminance value (Y_(3D) _(—) _(COMB) 646)may be written asY _(3D) _(—) _(COMB) =K _(—)3dcomb*Y _(—)3dcomb_(—) p+(1−K _(—)3dcomb)*Y_(—)3dcomb_(—) n.

In some embodiments, an overall luminance (Y_(MIX)) is based on mixingand/or selecting among a spatial luminance (Y_(2D) _(—) _(COMB)), atemporal luminance (Y_(3D) _(—) _(COMB)), and a notch luminance(Y_(NOTCH)). In a first embodiment, these three luminance values may bescaled and summed. Alternatively, a pair of these three luminance valuesmay be scaled and summed. Next, a selection may be made between theresulting scaled and summed luminance values and the third luminancevalue. For example, the spatial luminance (Y_(2D) _(—) _(COMB)) andtemporal luminance (Y_(2D) _(—) _(COMB)) may be mixed first as describedbelow with reference to FIG. 19. Next, a selection may be made betweenthis resulting luminance (Y_(COMB)) and the notch filtered luminance(Y_(NOTCH)).

In an alternative embodiment, the notch luminance (Y_(NOTCH)) andspatial luminance (Y_(2D) _(—) _(COMB)) are each scaled and the scaledvalued summed to produce an intermediate spatial luminance (Y_(2D)).Next, a selection is made between the intermediate spatial luminance(Y_(2D)) and the temporal luminance (Y_(3D) _(—) _(COMB)). Alternativelyto a selection between the intermediate spatial luminance (Y_(2D)) andthe temporal luminance (Y_(3D) _(—) _(COMB)), a weighting between thesevalues may be used. For example, if a pixel is determined to containrelatively still data as compared to a first threshold, then thetemporal luminance (Y_(3D) _(—) _(COMB)) may be selected. If the pixelis determined to contain relatively moving data relative to a secondthreshold, then the intermediate spatial luminance (Y_(2D)) may beselected. If the pixel is determined to be somewhere in between stilland moving (in between the first and second thresholds), then thetemporal luminance (Y_(3D) _(—) _(COMB)) and the intermediate spatialluminance (Y_(2D)) may be averaged to generate the overall luminance(Y_(MIX)).

FIGS. 19A and 19B show circuitry for mixing luminance values, inaccordance with the present invention. FIG. 19A shows circuitry formixing an intermediate spatial luminance value (Y_(2D)), describedfurther below, with a temporal luminance value (Y_(3D) _(—) _(COMB)).First, the spatial luminance value (Y_(2D)) is scaled by a scaler 705 bwith a first coefficent (a₁=K_y_mix). Next, the temporal luminance value(Y_(3D) _(—) _(COMB)) is scaled 706 b by a second coefficent (a₂=1−a₁).The results are summed with summer 707 b to produce an overall mixedluminance value (Y_(MIX)).

FIG. 19B shows circuitry for mixing spatial luminance values (Y_(NOTCH),Y_(2D) _(—) _(COMB)_up and Y_(2D) _(—) _(COMB) _(—) _(DOWN)) to producean intermediate spatial luminance value (Y_(2D) 709). The notchedfiltered luminance value (Y_(NOTCH)) and the spatial luminance value(Y_(—2D) _(—) _(COMB) _(—) _(up)) are scaled by scalers 704 a and 705 awith coefficients (a₁₁=1−K_up and a₁₂=K_up, respectively), then summedwith summer 707 a to produce a first intermediate spatial luminancevalue (Y_(2D) _(—) _(UP) 629 a). Similarly, the notched filteredluminance value (Y_(NOTCH)) and the spatial luminance value (Y_(2D) _(—)_(COMB) _(—) _(DOWN)) are scaled by scalers 704 b and 705 b withcoefficients (a₂₁=K_down and a₂₂=1−K_down, respectively), then summedwith summer 707 b to produce a second intermediate spatial luminancevalue (Y_(2D) _(—) _(DOWN) 629 b). The first intermediate spatialluminance value (Y_(2D) _(—) _(up) 629 a) and the second intermediatespatial luminance value (Y_(2D) _(—) _(DOWN) 629 b) are scaled byscalers 705 and 706 using coefficients (1−K_dirdown and K_dirdown,respectively), then summed with summer 707 to produce an intermediatespatial luminance value (Y_(2D)).

FIGS. 20A, 20B, 21, 22A, 22B and 23 show circuitry for computing spatialand temporal chrominance values, in accordance with the presentinvention. FIGS. 20A and 20B show example circuitry to compute spatialchrominance for a NTSC broadcasting system. FIGS. 21A and 21B showexample circuitry to compute spatial chrominance for a PAL broadcastingsystem. FIGS. 22 and 23 show example circuitry to compute temporalchrominance.

In FIG. 20A, circuitry computes a first spatial chrominance value(U_(2D) _(—) _(COMB)) based on raw chroma values U′ (u_p 631u, u_c 632u& u_n 633u) of a current field 602 from chroma demodulator 400 for theNTSC broadcasting system. A first raw chroma value u_p 631u is datagenerated at a center pixel c in a previous line 611. A second rawchroma value u_c 632u is data generated at a center pixel c in a currentline 612. A third raw chroma value u_p 633u is data generated at acenter pixel c in a next line 613. Values from the previous, current andnext lines may be buffered in memory. These chroma values are firstaveraged by summers 641u and 643u and then mixed by mixer 645u toproduce the first spatial chrominance value (U_(2D) _(—) _(COMB)). Thesecomputations may be implemented in various equivalent fashions with orwithout intermediate values. For example, the first spatial chrominancevalue may be written as:U _(2D) _(—) _(COMB)=(1−Kc_dirdw)*U _(—)2dcomb_(—) p+Kc_dirdw*U_(—)2dcomb_(—) n,

where a first chroma U value is the intermediate value:U _(—)2dcomb_(—) p=(u _(—) c+u _(—) p)/2,

where a second chroma U value is the intermediate value:U _(—)2dcomb_(—) n=(u _(—) c+u _(—) n)/2.

Similarly in FIG. 20B, circuitry computes a second spatial chrominancevalue (V_(2D) _(—) _(COMB)) based on raw chroma values V′ (v_p 631v, v_c632v & v_n 633 v) of a current field 602 from chroma demodulator 400. Afirst raw chroma value v_p 631v is data generated at the center pixel cin the previous line 611. A second raw chroma value v_c 632v is datagenerated at the center pixel c in the current line 612. A third rawchroma value v_p 633v is data generated at the center pixel c in thenext line 613. These chroma values are first averaged by summers 641vand 643v and then mixed by mixer 645v to produce the second spatialchrominance value (V_(2D) _(—) _(COMB)). In one example, the firstspatial chrominance value may be written as:V _(2D) _(—) _(COMB)=(1−Kc ₁₃ dirdw)*V _(—)2dcomb_(—) p+Kc_dirdw*V_(—)2dcomb_(—) n,

where a first chroma V value is the intermediate value:V _(—)2dcomb_(—) p=(v _(—) c+v _(—) p)/2,

where a second chroma V value is the intermediate value:V _(—)2dcomb_(—) n=(v _(—) c+v _(—) n)/2.

FIGS. 20A and 20B show different functional blocks (e.g., summers 641u &641v or mixers 645u & 645v) performing the identical functions exceptwith different input signals. These functions may be performed bydifferent hardware or software (as shown) or the functions may bepeformed sequencially by the same hardware.

In FIG. 21A, circuitry computes a first spatial chrominance value U_(2D)_(—) _(COMB) 669 based on raw chroma values U′ (including u_nu, u_nd,u_p, u_c & u_n) of a current field 602 from chroma demodulator 400 forthe PAL broadcasting system. Note that in a PAL broadcasting system,neighboring pixel may be used to further refine the first spatialchrominance value. Value u_nu represents a raw chroma value fromneighboring pixal u (“up”) as described in FIG. 12B. Value u_ndrepresents a raw chroma value from neighboring pixal d (“down”). Valuesu_p, u_c and u_n are the same as those above with reference to FIG. 20A.The first spatial chrominance value for the PAL system may be similarlywritten as:U _(2D) _(—) _(COMB)=(1−Kc_dirdw)*(U _(—)2dcomb_(—) p+U _(—)2dcomb_(—)nb)/2+Kc_dirdw* (U _(—)2dcomb_(—) n+U _(—)2dcomb_(—) nb)/2,

where U_(—)2dcomb_p=(u_c+u_p)/2,

where U_(—)2dcomb_nb=(u_nu+u_nd)/2, and

where U_(—)2dcomb_n=(u_c+u_n)/2.

In FIG. 21B, circuitry computes a second spatial chrominance valueV_(2D) _(—) _(COMB) 689 based on raw chroma values V′ (including v_nu,v_nd, v_p, v_c & v_n) of a current field 602 from chroma demodulator 400for the PAL broadcasting system. Value v_nu represents a raw chromavalue from neighboring pixal u (“up”) as described in FIG. 12B. Valuev_nd represents a raw chroma value from neighboring pixal d (“down”).Values v_p, v_c and v_n are the same as those above with reference toFIG. 20B. The second spatial chrominance value for the PAL system may bewritten as:V _(2D) _(—) _(COMB)=(1−Kc_dirdw)*(V _(—)2dcomb_(—) p+V _(—)2dcomb_(—)nb)/2+Kc_dirdw* (V _(—)2dcomb_(—) n+V _(—)2dcomb_(—) nb)/2,

where V_(—)2dcomb_p=(v_c+v_p)/2,

where V_(—)2dcomb_nb=(v_nu+v_nd)/2, and

where V_(—)2dcomb_n=(v_c+v_n)/2.

FIGS. 22 and 23 are shown to compute temporal chrominance values byreusing the circuitry of FIGS. 20A and 20B. Alternatively, the summersand mixers may be separate and dedicated to the functions of computingthe temporal chrominance values. In FIG. 22, circuitry computes a firsttemporal chrominance value based on a current raw chroma values U′(u_c_p, u_c_c & u_c_n) where u_c_p is a value from a current line and aprevious field 601, where u_c_c_n is a value from a current line and acurrent field 602 and where u_c_n is a value from a current line and anext field 603. The first temporal chrominance value may be written as:U _(3D) _(—) _(COMB) =U _(—)3dcomb_(—) p*Kmix_pre+U _(—)3dcomb_(—)n*(1−Kmix_pre)

where a first temporal chroma U value (a previous value p) is anintermediate value:U _(—)3dcomb_(—) p=(u _(—) c _(—) c+u _(—) c _(—) p)/2,

where a second temporal chroma U value (a next value n) is anintermediate value:U _(—)3dcomb_(—) n=(u _(—) c _(—) c+u _(—) c _(—) n)/2.

In FIG. 23, circuitry computes a second temporal chrominance value basedon a current raw chroma values V′ (v_c_p, v_c_c & v_c_n) where v_c_p isa value from a current line and a previous field 601, where v_c_c is avalue from a current line and a current field 602 and where v_c_n is avalue from a current line and a next field 603. The second temporalchrominance value may be written as:V _(3D) _(—) _(COMB) =V _(—)3dcomb_(—) p*Kmix_pre+V _(—)3dcomb_(—)n*(1−Kmix_pre)

where a first temporal chroma V value (a previous value p) is anintermediate value:V _(—)3dcomb_(—) p=(v _(—) c _(—) c+v _(—) c _(—) p)/2,

where a second temporal chroma V value (a next value n) is anintermediate value:V _(—)3dcomb_(—) n=(v _(—) c _(—) c+v _(—) c _(—) n)/2.

FIGS. 24 and 25 show circuitry for mixing spatial and temporal luminanceand chrominance values, in accordance with the present invention.

In FIG. 24, circuitry computes a final mixed luminance (Y_(MIX) 709), afirst mixed chrominance (U_(MIX) 719), a second mixed chrominance(U_(MIX) 729). The mixed luminance (Y_(MIX) 709) is a function of threeinput signals: a notched filtered luminance (Y_(NOTCH)), a luminance(Y_(2D) _(—) _(COMB) 729) and a temporal luminance (Y_(3D) _(—) _(COMB)749). The input signals are weighted by respective amplifiers 704, 705and 706 by values a₁, a₂ and a₃, which may typically be values rangingfrom zero to one. In some embodiments, the sum of a₁, a₂ and a₃ isunity. The first mixed chrominance (U_(MIX) 719) is a function of threeinput signals: a corrected chrominance (U′), a first spatial chrominance(U_(2D) _(—) _(COMB) 669) and a first temporal chrominance (U_(3D) _(—)_(COMB) 679). The input signals are weighted by respective amplifiers714, 715 and 716 by values b₁ (K_u_(—)2d), b₂ (1−b₁=1−K_u_(—)2d) and b₃,which may typically be values ranging from zero to one, and in someembodiments sum to unity. The second mixed chrominance (V_(MIX) 729) isa function of three input signals: a corrected chrominance (V′), asecond spatial chrominance (V_(2D) _(—) _(COMB) 689) and a secondtemporal chrominance (V_(3D) _(—) _(COMB) 699). The input signals areweighted by respective amplifiers 724, 725 and 726 by values c₁(K_v_(—)2d), c₂ (1−c₁=1−K_v_(—)2d)and c₃, which may typically be valuesranging from zero to one, and in some embodiments sum to unity.

FIG. 25 shows an alternative embodiment for circuitry 700 that computesa mixed luminance (Y_(MIX) 709), a first mixed chrominance (U_(MIX)719), a second mixed chrominance (U_(MIX) 729) based on spatial andtemporal luminance and chrominance values.

The mixed luminance (Y_(MIX) 709) is shown as a function of two inputsignals. The first of the two input signals is a spatial luminance(Y_(2D) 729 a), which is itself a function of a notched filteredluminance (Y_(NOTCH)) and a luminance (Y_(2D) _(—) _(COMB) 729) as shownpreviously. The second of two input signals is a temporal luminance(Y_(3D) _(—) _(COMB) 749). The input signals are weighted by amplifiers705 and 706 with values a₁ and a₂, respectively, which may typically bevalues ranging from zero to one. The weighted signals are summed bysummer 707 to produce the mixed luminance (Y_(MIX) 709). A mixingcoefficient K (e.g., where K=a₁=1−a₂ and where K=[0.0 to 1.0]) may bedetermined to allow a full degree of balancing spatial and temporalluminance values. In some embodiments, the sum of a₁ and a₂ is unity.For example, during operation if a spatial luminance is favored becausetemporal variations are too great, a₁ may be one and a₂ may be zero.Alternatively, if temporal variations are reliable so a temporalluminance is heavily favored, a₁ may be zero and a₂ may be one.Alternatively, if temporal and spatial variables are expected to befairly to marginally reliable, an even balance of spatial and temporalmay be provided with a₁=0.5 and a₂=0.5.

The first mixed chrominance (U_(MIX) 719) is shown as a function ofthree input signals: a first corrected chrominance (U′), a first spatialchrominance (U_(2D) _(—) _(COMB) 669) and a first temporal chrominance(U_(3D) _(—) _(COMB) 679). The spatial input signals, U′ and U_(2D) _(—)_(COMB) 669, are weighted by amplifiers 714 a and 715 a, respectively.The weighted signals are summed by summer 717 a to produce a firstintermediate spatial chrominance (U_(2D) 669a). The first intermediatespatial chrominance (U_(2D) 669a) along with the first temporalchrominance (U_(3D) _(—) _(COMB) 679) are inputs to a multiplexer 715with a stillness factor as a control signal. The Stillness factor iseither zero (Stillness=FALSE) if the pixel is considered to be a movingpixel and a one (Stillness=TRUE) if the pixel is considered to be astill pixel. A pixel is still if there is little to no change from fieldto field for that pixel position and is moving otherwise. Therefore, themultiplexer 715 acts to select the first intermediate spatialchrominance (U_(2D) 669 a) if the pixel position is determined to bemoving (Stillness=FALSE) and to select the first temporal chrominance(U_(3D) _(—) _(COMB) 679) if the pixel position is determined to bestill (Stillness=TRUE). The multiplexer 715 provided the selected signalas first mixed chrominance (U_(MIX) 719) at its output port.

Similarly, the first mixed chrominance (V_(MIX) 729) is shown as afunction of three input signals: a second corrected chrominance (V′), asecond spatial chrominance (V_(2D) _(—) _(COMB) 689) and a secondtemporal chrominance (V_(3D) _(—) _(COMB) 699). The spatial inputsignals, V′ and V_(2D) _(—) _(COMB) 689, are weighted by amplifiers 724a and 725 a, respectively. The weighted signals are summed by summer 727a to produce a second intermediate spatial chrominance (V_(2D) 689 a).The second intermediate spatial chrominance (V_(2D) 689a) along with thesecond temporal chrominance (V_(3D) _(—) _(COMB) 699) are inputs to amultiplexer 725 with the stillness factor, described above, as a controlsignal. Again, the multiplexer 725 acts to select the secondintermediate spatial chrominance (V_(2D) 689a) if the pixel position isdetermined to be moving and the second temporal chrominance (V_(3D) _(—)_(COMB) 699) if still. The multiplexer 725 provided the selected signalas second mixed chrominance (V_(MIX) 729) at its output port.

The figures provided are merely representational and may not be drawn toscale. Certain proportions thereof may be exaggerated, while others maybe minimized. The figures are intended to illustrate variousimplementations of the invention that can be understood andappropriately carried out by those of ordinary skill in the art.Therefore, it should be understood that the invention can be practicedwith modification and alteration within the spirit and scope of theclaims. The description is not intended to be exhaustive or to limit theinvention to the precise form disclosed. Furthermore, it should beunderstood that the invention can be practiced with modification andalteration.

1. A digital video resampler and decoder comprising: an input buffercoupled to a source of a video signal having horizontal synchronizationboundaries, the video signal being oversampled by at least two times; ahorizontal synchronization detector coupled to the input buffer, thehorizontal synchronization detector to detect horizontal synchronizationboundaries in the video signal; a counter coupled to the horizontalsynchronization detector, the counter to count a total number of samplesbetween a pair of the detected horizontal synchronization boundaries; acomparator coupled to the counter, the comparator to compare the countednumber of samples to a reference count; a sample corrector coupled tothe input buffer, wherein the sample corrector is operable to modify ablock of samples based on a result from the comparator to generate amodified block of samples; an output buffer coupled to the samplecorrector to hold the modified block of samples; and a comb filtercoupled to the output buffer, the comb filter to generating a first andsecond three-dimensional color values (U.sub.3D & V.sub.3D) based on themodified block of samples.
 2. The digital video resampler and decoder ofclaim 1, wherein the reference count is within eight of
 3432. 3. Thedigital video resampler and decoder of claim 1, wherein the correctorcomprises: an interpolator coupled to the input buffer, the interpolatorto modify a phase of the block of samples; and a resampler coupled tothe input buffer, the resampler to transform the counted number ofsamples to be a standard count different from the counted number ofsamples.
 4. The digital video resampler and decoder of claim 1, whereinthe corrector comprises: an interpolator coupled to the input buffer,the interpolator to modify a phase of the block of samples; an expandercoupled to the input buffer, the expander to transform the block ofsamples to be an increase number of samples; and a compressor coupled tothe input buffer, the compressor to transform the block of samples to bea decrease number of samples.
 5. A method of digital video decoding, themethod comprising: buffering samples of a video signal; detectinghorizontal synchronization boundaries in the video signal; counting atotal number of samples between a pair of the horizontal synchronizationboundaries; comparing the total counted number of samples to a referencecount; correcting samples in the buffer based on the act of comparingthe counted number of samples to the reference count to generate amodified block of samples; rebuilding a horizontal synchronizationsignal; generating a first three-dimensional color value (U.sub.3D)based on the modified block of samples; and generating a secondthree-dimensional color value (V.sub.3D) based on the modified block ofsamples.
 6. The method of claim 5, further comprising oversampling avideo signal by at least two times to generate the samples.
 7. Themethod of claim 5, wherein the act of detecting the horizontalsynchronization boundaries comprises filtering the samples.
 8. Themethod of claim 7, wherein the act of filtering the samples comprisesapplying a convolution and an integration.
 9. The method of claim 5,wherein the act of detecting the horizontal synchronization boundariescomprises: detecting a first horizontal synchronization pulse from thesamples signal; and detecting a second horizontal synchronization pulsefrom the samples signal.
 10. The method of claim 5, wherein thereference count is within eight of
 3432. 11. The method of claim 5,wherein the act of correcting samples comprises: interpolating thebuffered samples when the counted number of samples is within a firstrange; compressing the buffered samples when the counted number ofsamples is larger than a first range; and expanding the buffered sampleswhen the counted number of samples is smaller than a first range. 12.The method of claim 5, wherein the act of correcting samples comprisesgenerating new samples between pairs of buffered samples byinterpolating based on the buffered samples and the counted number ofsamples.
 13. The method of claim 5, wherein the act of correctingsamples comprises resampling the samples.
 14. The method of claim 13,wherein the act of resampling the samples comprises expanding thesamples to generate additional samples.
 15. The method of claim 13,wherein the act of resampling the samples comprises compressing thesamples to reduce samples.
 16. The method of claim 5, wherein the act ofgenerating the first and second three-dimensional color values (U.sub.3D& V.sub.3D) comprises applying a comb filter.
 17. The method of claim 5,further comprising: generating a luminance value (Y.sub.COMB).
 18. Adigital video resampler and decoder comprising: means for bufferingsamples of a video signal; means for detecting horizontalsynchronization boundaries in the video signal; means for counting atotal number of samples between a pair of the horizontal synchronizationboundaries; means for comparing the counted total number of samples to areference count; means for correcting samples in the buffer based on theact of comparing the counted number of samples to the reference count togenerate a modified block of samples; means for rebuilding a horizontalsynchronization signal; means for generating a first three-dimensionalcolor value (U.sub.3D) based on the modified block of samples; and meansfor generating a second three-dimensional color value (V.sub.3D) basedon the modified block of samples.